Automatic testing device for signal evaluation

ABSTRACT

A phase noise measurement system including a low noise programmable synthesizer and a receiver/down converter is provided. The low noise synthesizer provides L-Band Signals which can selectively exhibit low noise close-in or low noise far out. The receiver down converter provides for absolute, additive, and down converted/direct/multiple phase noise measurement.

FIELD OF THE INVENTION

The present invention relates to the field of automatic test equipmentfor testing electronic signals, and more particularly, to automatic testequipment for analyzing the noise component of an electronic signal.

BACKGROUND OF THE INVENTION

Automatic test equipment for testing the performance of communicationssystems, radar systems, and other signal producing devices are known. Inthis regard, it is often necessary to evaluate the spectral purity of asignal produced by a unit under test (UUT) in order to determine if theUUT is operating within the manufacturer's specifications. Specifically,a manufacturer or end user may specify the maximum phase noise which maybe present on a signal produced by the UUT. The phase noise of a signalis a measure of the random phase instability of the signal.

The phase noise of a UUT can be measured in a variety of ways. Forexample, the output signal of the UUT can be applied directly into theinput of a spectrum analyzer which will display the power spectraldensity of the signal and the phase noise will be visible in the displayas random noise power in the spectral plot. Alternatively, the phasenoise can be measured using a second signal source as a reference. Thesecond signal source outputs a signal which is identical to or betterthan the expected UUT signal, but in phase quadrature (if phasemodulation noise is being tested) to the UUT signal, i.e. the secondsignal source is at the same frequency as the UUT signal, but is phaseshifted by 90 degrees. The UUT and the second signal source are inputinto a mixer, and, since the two signals have the same carrierfrequency, the signals cancel each other out, leaving a signalcomprising the combined phase noise of the UUT and the second signalsource. In addition, the phase noise may be measured using the DownConverter/Multiple Direct Spectrum Measurement Technique, which isdescribed in U.S. Pat. Nos. 5,337,014 and 5,179,344, the specificationsof which are hereby incorporated by reference.

Conventionally, when a UUT signal is analyzed, the technician uses avariety of discrete components including a programmable down converterfor translating the input signal into a lower, and more easily analyzed,frequency; a narrow FM tunable synthesizer for generating a referencesignal; and a separate spectrum analyzer. Since each of these componentshas their own unique programming requirements, significant time andeffort is often spent programming and integrating these discretecomponents into an effective phase noise measurement system.

SUMMARY OF THE INVENTION

In accordance with the present invention, an integrated phase noisemeasurement system is provided which includes a low noise synthesizermodule, a receiver/downconverter module, a controller, a digitizer, anda spectrum analyzer. The low noise synthesizer andreceiver/downconverter may be used separately, or in combination withone another.

The low noise synthesizer module may be used in a phase noisemeasurement system to produce, for example, spectrally pure L-Bandsignals. In phase noise measurement systems, it is important that thesynthesizer module, which provides the reference signal to which the UUTis compared, produce an extremely low noise signal. This is importantbecause the phase noise measurement system will be unable to accuratelymeasure noise in the signal produced by the UUT which is below the noiselevel of the synthesizer. As a result, the noise level of thesynthesizer sets a "noise floor" below which noise measurements cannotbe made. In addition, it is sometimes desirable to measure the noise ofa signal "far out" as well as "close-in" from the carrier frequency ofthe UUT. Conventional low noise synthesizers, however, are normallyunable to provide a reference signal which exhibits low noise both "farout" as well as "close-in" from the carrier frequency.

In accordance with the present invention, a low noise synthesizer isprovided which produces an output signal with a low (phase andamplitude) noise characteristic close in to the carrier, as well as lownoise far out from the carrier, by utilizing a low noise oscillatorcoupled to a comb generator to provide a signal with low noise close in,and a signal acoustic wave oscillator to provide a signal with low noisefar out. The low noise synthesizer includes a low noise crystaloscillator for producing a signal having a frequency (preferably 120MHZ), and a surface acoustic wave oscillator for producing a signalhaving a second frequency (preferably 960 MHz). A comb generator iscoupled to an output of the crystal oscillator, and a bandpass filter iscoupled to the output of the comb generator. The bandpass filter has apassband which includes the second frequency (and preferably has apassband centered at 960 MHz). A frequency dividing component is coupledto the low noise crystal oscillator to selectively produce one of aplurality of offset frequencies, each of the plurality of offsetfrequencies being in a first frequency range (which is preferably from10-40 MHz). A mixer has a first input coupled to an output of thefrequency dividing component, and has a second input which isselectively coupled to either the output of the surface acoustic waveoscillator or the output of the bandpass filter. A tunable bandpassfilter is coupled to an output of the mixer, and is selectively tuned toa passband which includes a sum (or difference) of the selected offsetfrequency and the second frequency. The output of the tunable bandpassfilter provides an output signal with low noise close in when the outputof the bandpass filter is coupled to the mixer, and an output signalwith low noise far out when the surface acoustic wave oscillator iscoupled to the mixer.

In accordance with the preferred embodiment of the low noisesynthesizer, an low noise oscillator produces a 120 MHZ reference signalwhich is multiplied to 960 MHz by a comb generator to provide a signalwith low noise close in, (e.g. within 400 KHZ of the carrier frequency)while a surface acoustic wave oscillator is utilized to produce a 960MHZ signal to provide a signal with low noise far out (e.g. >carrierfrequency+400 KHZ, <carrier frequency-400 KHZ) but relatively high noisewithin 400 KHZ of the carrier frequency. In accordance with thispreferred construction, a noise component of less than 100 dBc isachieved at 100 Hz, <120 dBc is achieved at 1 KHz, <130 dBc is achievedat 10 KHZ, <140 dBc is achieved at 100 KHZ, and <160 dBc is achievedat >=400 KHz.

The receiver/downconverter in accordance with the present invention ispreferably used in conjunction with the low noise synthesizer describedabove, but may also be used with conventional synthesizers. Inaccordance with the present invention, the receiver/downconverter can beused to perform absolute phase noise measurement, additive phase noisemeasurement, and down converter/multiple direct (DMD) phase noisemeasurement.

The receiver/downconverter includes a UUT input, a synthesizer input, anoutput, a first mixer, a second mixer, a delay element coupled to afirst phase shifter, a phase locked loop (PLL) coupled to a second phaseshifter, a first bandpass filter, a second band pass filter, a first lowpass filter, a second low pass filter, a comb generator, and anamplifier. The synthesizer input is coupled to one input of the firstmixer. The UUT input is coupled directly through to the other input ofthe first mixer for absolute phase noise measurement, and for DMD phasenoise measurement. For an additive phase noise measurement, the UUTinput is coupled through the delay element and first phase shifterbefore being applied to the other input of the first mixer. The outputof the first mixer is selectively coupled to one of the first bandpassfilter, the second bandpass filter, and the first low pass filter. Theoutput of the second bandpass filter is coupled to the comb generator,and the output of the first bandpass filter is coupled to the PLL and toone input of the second mixer. The PLL is coupled to the second phaseshifter, which, in turn, is coupled to the second input of the secondmixer. The output of the second mixer is coupled to the second low passfilter. Finally, the outputs of the first low pass filter, the secondlowpass filter, and the comb filter, are selectively coupled to theamplifier. The amplifier is coupled to the output of thereceiver/downconverter.

In order to perform an absolute phase noise measurement with thereceiver/downconverter, a synthesizer signal is generated by thesynthesizer which is offset by a first offset frequency (preferably 10MHz) from the expected UUT signal frequency. The synthesizer signal andthe UUT signal are applied to the first mixer to generate an IF signal(preferably 10 MHz IF), which is then coupled through to the firstbandpass filter (10 MHz BP for 10 MHz IF signal) to isolate the IFsignal. The IF signal is then coupled to both i) the input of the PLL,and ii) one input of the second mixer. The output of the PLL is thenpassed through the second phase shifter prior to being applied to theother input of the second mixer. The PLL-Phase shifter circuit maintainsphase quadrature between the two inputs of the second mixer. Since thetwo inputs to the second mixer have the same frequency, the mixer willoutput a 0 MHZ difference signal to the second low pass filter(preferably 1.9 MHz low pass). The signal output from the second lowpass filter, which comprises the absolute noise signal, is then passedthrough to the amplifier, and then output to a spectrum analyzer via theoutput of the receiver/downconverter.

In order to perform an additive phase noise measurement with thereceiver/downconverter, a synthesizer signal is generated by thesynthesizer which is equal in frequency to the expected UUT signal. TheUUT signal is passed through the delay line and the first phase shifterbefore being applied to one input of the first mixer. The first phaseshifter maintains phase quadrature between the two inputs to the firstmixer. Since the signals input to the first mixer are of equalfrequency, the first mixer will output a 0 MHz difference signal to thefirst low pass filter (preferably 1.9 MHz low pass). The signal outputfrom the second low pass filter, which comprises the additive phasenoise signal, is then passed through to the amplifier, and output to thespectrum analyzer via the output of the receiver/downconverter.

In order to use the receiver/downconverter to down convert and multiplythe UUT signal to microwave frequencies (DMD phase noise measurement)which can be directly measured on a spectrum analyzer as a double sideband signal, a synthesizer signal is generated by the synthesizer whichis offset by a second offset frequency (preferably 100 MHz) from theexpected UUT signal frequency. The synthesizer signal and the UUT signalare applied to the first mixer to generate a second IF signal(preferably 100 MHz IF), which is then coupled through the thirdbandpass filter (preferably 100 MHz), and into the comb generator. Asone of ordinary skill in the art will appreciate, the comb generatorproduces a spectrum of signals which are multiples of the inputfrequency signal. The output of the comb generator is then passedthrough the amplifier and on to the spectrum analyzer. The user can thenchoose which multiple of the second IF frequency he or she wishes toview. Preferably, a signal in the 4-6 Ghz range is chosen because thatis the most effective range of most spectrum analyzers.

In accordance with another embodiment of the present invention, thereceiver/downconverter and the low noise synthesizer are incorporatedinto an integrated phase noise measurement system including a spectrumanalyzer, a digitizer, and a controller. The controller may include, forexample, a computer, a display screen, and a keyboard.

In order to measure absolute phase noise, an operator specifies thecarrier frequency of the UUT, and indicates whether the signal to bemeasured is AM (amplitude modulated) or PM (phase modulated). Thisinformation is input to the controller via the keyboard, input program,or other input device. The controller automatically configures thereceiver/downconverter to receive a 10 MHz IF. This 10 MHz signal ismixed with the PLL 10 MHz signal and a residual error frequency ismeasured on the digitizer. After correcting this residual error byautomatically reprogramming the frequency difference to account for thiserror, AM or PM noise is measured. At this point, when the PLL & 10 MHzIF signals are identical in frequency, the phase locked PLL and 10 MHzIF signals are phase detected and the second phase shifter is rotatedthrough zero crossings and peak amplitudes to establish the beat noteamplitude and to establish a quadrature setting for PM or peak settingfor A. This determinant also provides the DC beat note level from thephase detector. If the measurement is being made on an AM (amplitudemodulated) signal, then the peak signal value is stored as the beat note(PM signals will have a constant amplitude). The beat note representsthe total RF power after phase detector conversion loss, whichrepresents the signal level that the noise spectrum is referenced to.The controller then switches the output of the amplifier of thereceiver/downconverter through to the spectrum analyzer. The powerspectral density and spurious is then measured by the spectrum analyzer.Then, in order to establish true power spectral density, the controllerapplies 55 dB, 3 dB tangential, 3 dB single sideband, and 2.5 dBFilter/log Fidelity corrections to the measured noise power. Finally,the controller rechecks the quadrature (for PM measurement) or peak (forAM measurement) setting to ensure that quadrature or peak was not lostduring the foregoing measurement.

In order to measure additive phase noise, an operator specifies thecarrier frequency of the UUT, and indicates whether the signal to bemeasured is AM (amplitude modulated) or PM (phase modulated). Thisinformation is input to the controller via the keyboard, input program,or other input device. The controller automatically configures thereceiver/downconverter to provide an additive phase noise measurement,and couples the output of the first phase shifter of the receiverdownconverter to the digitizer. The controller then rotates the firstphase shifter until quadrature is established and stores the phaseshifter value. The output from the first phase shifter is then digitizedby the digitizer and transmitted to the controller for processing. Thecontroller processes the digitized data to determine the beat note ofthe UUT signal. If the measurement is being made on an AM (amplitudemodulated) signal, then first phase shifter is rotated until peak isestablished, and the peak signal value is stored as the beat note (PMsignals will have a constant amplitude). The controller then switchesthe output of the amplifier of the receiver/downconverter through to thespectrum analyzer. The power spectral density and spurious is thenmeasured by the spectrum analyzer. Then, in order to establish truepower spectral density, the controller applies 55 dB, 3 dB tangential, 3dB single sideband, and 2.5 dB Filter/log Fidelity corrections to themeasured phase power. Finally, the controller rechecks the quadrature orpeak setting to insure that quadrature or peak was not lost during theforegoing measurement.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a preferred embodiment of a phase noise measurement systemin accordance with the present invention.

FIG. 2 shows a preferred embodiment of an stable local oscillatorsynthesizer in accordance with the present invention.

FIG. 3 shows a preferred embodiment of a receiver/downconverter inaccordance with the present invention.

FIG. 4 shows a preferred embodiment of a flowchart for performing anabsolute phase noise measurement in accordance with the presentinvention.

FIG. 5 shows a preferred embodiment of a flowchart for performing anadditive phase noise measurement in accordance with the presentinvention.

FIG. 6 shows a preferred embodiment of a flow chart for DMD phase noisemeasurement in accordance with the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 shows a phase noise measurement system in accordance with anembodiment of the invention. A Unit Under Test 1000 is coupled to aphase noise measurement system 2000 which includes a stable localoscillator synthesizer 1, a receiver/downconverter 2, a digitizer 3, aspectrum analyzer 4, and a computer 5 including a display screen 6,keyboard 7, and mouse 8.

FIG. 2 shows the stable local oscillator synthesizer 1 (STALO) in moredetail. The synthesizer is a programmable module which selectivelyproduces L-Band signals and S-Band signals. In accordance with thisembodiment, the L-Band signal can selectively provide a low noise signal"close in", e.g within ˜400 KHz of the output signal, or low noise farout, e.g. greater than ˜400 KHz from the output signal. In accordancewith this construction, a noise component of less than 100 dBc isachieved at 100 Hz, <120 dBc is achieved at 1 KHz, <130 dBc is achievedat 10 Khz, <140 dBc is achieved at 100 Khz, and <160 dBc is achievedat >=400 KHz. In accordance with this aspect of the invention, anoscillator 10 produces a 120 MHz reference signal which is multiplied bya comb generator 60 to produce a 960 MHz signal with low noise in the960 MHz +/-400 KHz range to provide a low noise signal close in. Incontrast, a SAW 100 (surface acoustic wave oscillator) is utilized toproduce a 960 MHz signal with low noise at frequencies far out(i.e. >960.4 MHz, <959.6 MHz), but relatively high noise close in (i.e.960 MHz+/-400 KHz).

Oscillator 10 produces a 120 MHz reference signal which is used togenerate L-Band and S-Band reference signals for use in measuring theabsolute phase noise of a UUT. The signal from the oscillator 10 ispassed through a low pass filter 20 (150 MHz cutoff) to remove highfrequency noise, and then through a power divider 30 which produces 120MHz signals at each of its outputs 1 & 2. From output 1 of the powerdivider 30, the 120 MHz signal is input into a programmable bias network270, having outputs 2, 3, and 4. The bias network 270 comprises avariety of amplifiers, dividers, and filters which are configured asknown in the art to provide a programmable output of 10-40 MHz at output2, a programmable output of 10-40 MHz at output 3, and a fixed output of60 MHz at output 4. The outputs at 2 and 3 are then passed throughrespective amplifiers and 39 MHz low pass filters (280 & 290, 480 & 490)to provide respective programmable output signals of 10-39 MHz.

When the synthesizer 1 is programmed to produce an L-Band signal withlow noise close in, the 120 MHZ output of power divider 30 passesthrough directional couple 40 and into a comb generator 60. As one ofordinary skill in the art will appreciate, a comb generator produces aspectrum of signals which are multiples of the input frequency signal.Consequently, the output of the comb 60 will comprise a plurality ofsignals having frequencies equal to multiples of 120 MHZ. Band passfilters 70, 80 isolate the 960 MHz signal portion of the comb generator60 output, and pass it through switch 150 to low pass filter 160 (100MHz) to remove high frequency noise. The 960 MHz signal then passesthrough switches 170 and 210 and into the L input of Mixer 400. Theprogrammable 10-39 MHz signal from output 3 of Bias Network 270 ispassed through tunable bandpass filter 580 (which is tuned to the valueof the signal from output 3) to remove low end and high end noise. Thesignal is then passed through power divider 510, whose output 2 isprovided to the R input of mixer 360. Finally, the 120 MHz output ofdirectional couple 40 passes through amplifier 240, switches 250, 340,350, and low pass filter 350 (150 MHz) to provide a 120 MHz signal tothe L input of Mixer 360. The Mixer 360 produces a sum signal (120MHz+(10-30 MHz)) and a difference signal (120 MHz-(10-30 MHz)) at itsoutput I. Therefore, if, for example, the bias network 270 is programmedto provide an output of 10.2 MHz, then the signals available at theoutput of Mixer M3 will be 109.8 MHz and 130.2 MHz. Either of thesesignals could be selected by tunable bandpass filter 370. Consequently,by properly programming the bias network 270, and the filters 800, 370,signals in the range from 90 MHz through 150 MHz may be selected forinput into the R input of Mixer 400 via tunable bandpass filter 370(passband width of 200 KHz), amplifier 380, and low pass filter 390. TheMixer 360 produces a sum signal ((90-150 MHz)+960) and a differencesignal (960 MHz-(90-150 MHz) ) at its output I. In this case, only thesum signal (1050 MHz to 1110 MHz) is passed through to the L-band Outputvia tunable Yig Filter 420, amplifier 430, power divider 220, and lowpass filter 230. The use of the low noise 120 MHz oscillator multipliedby the comb generator to 960 MHz provides low signal noise within 400KHz of the 960 MHz signal. However, due to the comb generatormultiplication, the signal noise far out has been increased. In order toprovide an increased L-Band range, the 960 MHz signal can be mixed atmixer 180 with the output of a 300 MHz oscillator 260 to create a 1260MHz signal which, in turn, is applied to a 1260 MHz bandpass filter 190.In general, the amplifiers in the circuits of FIGS. 2 and 3 are providedto counteract gain losses caused by the other components, to providebetter reverse isolation, and to provide gain to levels >+10 dBm for theL-Band output signals.

In order to produce an L-Band signal with low noise far out, a SAWoscillator (surface acoustic wave) 100 is utilized to produce a 960 MHzsignal with low noise far out (more than 400 KHz from the carrierfrequency). This 960 MHz signal is passed through switch 120, amplifier130, switches 140 and 150, low pass filter 160, and switches 170, 210,and finally is fed into mixer 400. The remainder of the circuit operatesin the manner described above for L-Band signal with low noise near in.A SAW 101 may be provided which produces a 1030 MHz signal in order toprovide a greater L-Band range.

In order to produce an S-Band signal, the output of power divider 220 isfed into a trippler which triples the L-Band signal into an S-Bandsignal. The bandpass filter 450 is set to pass only the 3 by 0 productoutput of the mixer.

The L-Band output of the synthesizer 1 is applied to Receiver/Downconverter 2 at input 12. The output of the UUT 1000 is supplied to input9 of the receiver/down converter 2. The Receiver/Down converter 2 can beused:

1) to perform absolute phase noise measurement of L Band and VHF by downconverting the UUT signal to a 10 MHz IF, passing the 10 MHZ IF signalthrough a PLL and phase shifter, rotating the phase shifter to establishquadrature (or peak for measurement) and passing the 10 MHZ IF PLL inputsignal and the 10 MHz IF PLL output signal through a mixer to obtain 0MHz difference signal, and outputting the 0 MHz difference signal to thespectrum analyzer for an absolute phase noise Measurement;

2) to perform additive phase noise measurement of S Band signals bypassing an S-Band signal from the UUT through a delay line and phaseshifter and into the R input of a mixer, applying an S-Band signal ofequal carrier frequency from the synthesizer to the L input of themixer, controlling the phase shifter to establish phase quadrature (orpeak for AM measurement), isolating the 0 MHz difference output signalof the mixer with a 1.9 MHz low pass filter, and applying the filteredsignal to the spectrum analyzer for an additive phase noise measurement;

3) to down convert and multiply the UUT signal to microwave frequencieswhich can be directly measured on a spectrum analyzer as a double sideband signal by downconverting the UUT signal to a 100 MHz IF,multiplying the 100 MHz IF to microwave frequencies using a combgenerator, and then outputting the multiplied signals to a spectrumanalyzer for DMD phase noise measurement.

The manner in which an absolute phase noise measurement is made will nowbe described with reference to FIGS. 3 and 4.

In order to perform an absolute phase noise measurement for an L-Bandsignal, a technician will connect the UUT output to input 9 and connectthe synthesizer 1 L-Band output 17 to receiver 2 input 12 (step 5010)and will specify 1) whether the absolute phase noise measurement will befor AM (amplitude modulation) or PM (phase modulation) absolute; and 2)the frequency of the UUT signal. This information is input to thecomputers via keyboard 7.

Once this information has been input, the computer 5 will set thesynthesizer 1 to output a signal which is 10 MHz higher (or lower) thanthe specified UUT signal frequency. In the receiver 2, the UUT signalwill pass from input 9 through switches 840, 860, 870, 910, 920, and 930into the R input of Mixer 700. The L-Band signal from the synthesizer 1,which is offset 10 MHz from the UUT signal, is applied to the L input ofMixer 700. The output I of the mixer 700 then passed through switches710, 720 and into 10 MHz bandpass filter 730. In this manner, the UUTsignal has been down converted to a 10 MHz IF.

The downconverted 10 MHzIF is simultaneously applied to (a) PLL1200which includes a VCXO oscillator 1211 and to (b) amplifier 1000 and theR input of mixer 980, such that the 10 MHz VCXO signal of the VCXOoscillator 1211 will phase lock to the signal applied to the PLL 1200 atwhatever phase stabilizes the loop. This VCXO signal is decoupled andapplied to a phase shifter 1201 and then through attenuator 950,amplifier 960, and into the L input of mixer 980. In this regardattenuator 950 reduces the signal gain to within the dynamic range ofamplifier 960. The digitizer is coupled to J12 of switch 1240 and thephase shifter is rotated through 180 degrees in order to establish peak(for AM), or quadrature (for PM) and the beat note of the canceledsignal is measured (any noise measured is PM or AM noise). In thismanner the 10 MHz down converted UUT signal has been canceled, leavingsignal noise in the range of 1 Hz to 1.9 MHz at the output of the filter990. The noise signal is then passed through switch 1000 and into a lownoise amplifier 1230 (which provides a programmable gain of 45 to 50dB). The low noise programmable amplifier 1230 is then programmed(nominal>50 dB gain) and the noise signal is applied to the spectrumanalyzer from output J11 of switch 1240. Since the amplifier 1230 has aeffective range of from 75 Hz to 5 MHz, it will not adversely affect thenoise signal. The gain provided by the amplifier 1230 is used toincrease the noise signal by a known factor (without significantdistortion) so that the noise signal is in the effective range of thespectrum analyzer. Power spectral density is then evaluated from 100 Hzto 2 MHz and the data is automatically evaluated and plotted foradditional operational evaluation.

Referring to FIG. 4, in order to measure the absolute phase noise, oncethe operator has verified that the PLL is phase locked, the phaseshifter 1201 is rotated as set forth above until quadrature (for PMmeasurement) or peak (for AM measurement) is established between theoutput signal at J4 and the input signal at J2 (FIG. 4, step 5030) Oncequadrature is established for PM measurement, the beat note is measuredon the digitizer (FIG. 4, step 5040). If the measurement is being madeon an AM (amplitude modulated) signal, then the peak signal value isstored as the beat note (PM signals will have a constant amplitude). Thebeat note represents the total RF power after phase detector conversionloss which represents the signal level that the noise spectrum isreferenced to. The output of amplifier 1230 is then switched through tothe spectrum analyzer via switch 1240 (FIG. 4, step 5050). The powerspectral density and spurious is then measured by the spectrum analyzer(FIG. 4, step 5060). Then, in order to establish true power spectraldensity, 55 dB, 3 dB tangential, 3 dB single sideband, and 2.5 dBFilter/log Fidelity is applied as set forth in the flow chart of FIG. 4(step 5070). Finally, the quadrature setting (for PM measurement) orpeak setting (for AM measurement) is rechecked to insure that thesetting was not lost during the foregoing measurement (FIG. 4, step5080).

Referring to FIGS. 3 through 5, in order to perform an additive phasenoise measurement, the S-band output of the synthesizer is coupled toinput 12 of the receiver 2, and the S-band output of the UUT is appliedto the input 9 of the receiver 2 (FIG. 5, step 6010). The signal inputat input 9 passes through switch 840, switch 860, 870, and then throughdelay lines 880, 890, (40 nanoseconds) through phase shifter 900,switches 910, 920, 930 and then into the R input of mixer 700. Mixer 700mixes the S-band output of synthesizer 1 with the delayed version of theS-band UUT signal, producing a difference output of 0 MHz at the outputof the mixer, which is isolated by 1.9 MHz low pass filter 750, passedthrough switches 760, 780, low noise amplifier 1230, and then outputthrough switch 1240 to either a digitizer or the spectrum analyzer.Delay lines 880, 890 serve to reduce the synthesizer phase noise whenthe phase shifter maintains quadrature between the inputs to the mixer700 so that the synthesizer phase noise rise is low enough to ensureadditive phase noise sensitivity.

Referring to FIG. 5, in order to measure the additive phase noise, theoutput of the mixer 700 is coupled through the amplifier 1230 and isswitched to the digitizer via switch 1240 for PM measurement. The phaseshifter 900 is rotated until quadrature is established with the S-Bandoutput of the synthesizer (Steps 6020, 6030). The phase shifter value isthen stored, and the beat note is measured by the digitizer. It shouldbe noted that although the frequency signal is canceled, a dc outputstill exists which is proportional to phase difference between inputsignals at the mixer 700 phase detector. Therefore, as the phase shifter700 is rotated, the peak to peak signal level is determined and is usedto evaluate beat note and to establish the quadrature versus peak phaseshifter setting. The beat note represents the total RF power after phasedetector conversion loss which represents the signal level that thenoise spectrum is referenced to. The output of amplifier 1230 is thenswitched through to the spectrum analyzer via switch 1240 (Step 6050).The power spectral density and spurious is then measured by the spectrumanalyzer (Step 6060). Then, in order to establish true power spectraldensity, 55 dB, 3 dB tangential, 3 dB single sideband, and 2.5 dBFilter/log Fidelity is applied as set forth in Step 6070. Finally, thequadrature setting is rechecked to insure that quadrature was not lostduring the foregoing measurement (Step 6080).

If the measurement is being made on an AM (amplitude modulated) signal,the same procedure is followed except that the phase shifter 1201 isrotated until peak is established, and the peak signal value is storedas the beat note (Step 6040).

In certain applications, the UUT 1000 produces its S-Band signal bypassing its L-Band signal through a UUT tripler (not shown). In suchapplications, the UUT tripler can be separately tested in the followingmanner. First, the S-band output of the synthesizer is coupled to input12 of the receiver 2, and the S-band output of the UUT is applied to theinput 9 of the receiver 2. Additive phase noise measurement is thenperformed in the same manner described above. Since both the UUT signaland the synthesizer signal originate from the L-Band output of thesynthesizer, the resulting noise measurement can be compared to theknown noise of the synthesizer 1 to determine the noise which has beenadded by the UUT tripler.

The noise of the synthesizer 1 is measured during a calibrationprocedure by applying output 1 of power divider 470 of synthesizer 1 toinput 12 of the receiver 2, applying output 2 of the power divider 470to input 9 of the receiver, and following the procedures for additivephase noise measurement which we described above. In this manner, thereceiver 2 mixes the S-band output of synthesizer 1 with a delayedversion of itself in order to produce an output signal comprising thenoise of the synthesizer 1. This noise can then be subtracted from thetotal noise measured during the additive phase noise measurementsdescribed above in order to determine the noise added by the UUT and/orUUT tripler respectively.

The procedure for the DMD direct measurement technique will now beexplained with reference to FIG. 6. In this mode the synthesizer 1 isprogramed to produce a signal which is 100 MHZ offset (higher or lower)from the expected UUT signal. The synthesizer output is applied to input9 of the receiver 2 and the UUT output is applied to input 12 of thereceiver (step 7106). The UUT signal at input 12 passes through switches840, 860, 870, 910, 920 and into the R input of mixer 700. Mixer 700mixes the UUT signal (at the R input) with the 100 MHZ offset signalfrom synthesizer 1 (at the L input) to create a 100 MHZ IF signal atoutput I of mixer 750. The 100 MHZ IF signal is passed through 100 MHZbandpass filter 740, through switch 810 and into comb generator 820(step 7200). As explained above, a comb generator generates a series ofpulse bands which have frequencies equal to multiple of the inputsignal. Therefore the comb generator 820 produces output signals at 100MHz, 200 MHz, 300 MHz . . . , 16 GHz, 1.1 GHz . . . et al. This signalgoes directly out to the spectrum analyzer for noise measurement as adouble side band signal. The operator can then choose which picket hewishes to work with. Preferably, the operator will choose a picket inthe 4-6 GHZ range because this is in the most effective range of thespectrum analyzer. In any case, the operator sets up the spectrumanalyzer to measure a preselected comb frequency, and thedouble-sideband noise power is measured at +/-100 Hz, +/-1 KHz, +/-10KHz, +/-100 KHz, +/-2 MHz (step 7300). The above procedure results in a20 Log N increase in noise power (i.e., the noise received by thespectrum analyzer is 20 Log N times the actual UUT noise) where N is theselected comb frequency. Thus, in step 7400, the 20 Log N increase isdetermined, and the noise display on the spectrum analyzer is modifiedto reflect the actual UUT noise. At step 7500, the system determineswhether all tests have passed, i.e., whether the noise measured in theabsolute phase noise measurement, the additive phase noise measurement,and DMD phase noise measurement is within a predetermined range. If alltests have not been passed, the operator is instructed to replace andrepair (R/R) the UUT (step 760).

What is claimed is:
 1. A receiver/downconverter, comprising:a UUT input;a synthesizer input; an output; a first bandpass filter; a second bandpass filter; a first low pass filter; a first phase shifter, the UUTinput being selectively coupled to an input of the phase shifter foradditive phase noise measurement; a first mixer, a first input of thefirst mixer being coupled to the synthesizer input, a second input ofthe first mixer being selectively coupled to UUT input for absolutephase noise measurement and DMD phase noise measurement, the secondinput being selectively coupled to an output of first phase shifter foradditive phase noise measurement, an output of the first mixer beingselectively coupled to a first bandpass filter for absolute phase noisemeasurement, the output of the first mixer being selectively coupled tothe second bandpass filter for DMD phase noise measurement, and theoutput of the first mixer being selectively coupled to the first lowpass filter for additive phase noise measurement; a second mixer havinga first input, a second input, and an output, an output of the firstbandpass filter being coupled to the first input of the second mixer; aphase locked loop coupled to a second phase shifter, an output of thefirst bandpass filter being coupled to the phase locked loop, an outputof the second phase shifter being coupled to the second input of thesecond mixer; a second low pass filter having an input coupled to theoutput of the second mixer; and a comb generator coupled to an output ofthe second bandpass filter, the outputs of the first low pass filter,the second lowpass filter, and the comb generator, being selectivelycoupled to the output of the receiver/downconverter.
 2. Thereceiver/downconverter according to claim 1, further comprising a delayelement, the delay element being coupled between the input of the firstphase shifter and the UUT input for additive phase noise measurement. 3.The receiver/downconverter according to claim 1, wherein the passband ofthe first bandpass filter is centered at 100 MHz.
 4. Thereceiver/downconverter according to claim 1, wherein the passband of thesecond bandpass filter is centered at 10 MHz.
 5. Thereceiver/downconverter according to claim 1, wherein the cutofffrequency of the first and second lowpass filters is 1.9 MHz.
 6. A phasenoise measurement system, comprising:a controller; a low noisesynthesizer for selectively generating L-Band and S-Band signals, thelow noise synthesizer being coupled to the controller, the controllercontrolling the low noise synthesizer to selectively generate signals inthe L-Band and S-Band ranges; a receiver/downconverter coupled to thecontroller, the receiver/downconverter includinga UUT input; asynthesizer input for coupling to an output of the low noisesynthesizer; an output; a first bandpass filter; a second band passfilter; a first low pass filter; a first phase shifter, the controllerselectively coupling the UUT input to an input of the phase shifter foradditive phase noise measurement; a first mixer, a first input of thefirst mixer being coupled to the synthesizer input, the controllerselectively coupling a second input of the first mixer to the UUT inputfor absolute phase noise measurement and DMD phase noise measurement,the controller selectively coupling the second input to an output offirst phase shifter for additive phase noise measurement, the controllerselectively coupling an output of the first mixer to a first bandpassfilter for absolute phase noise measurement, the controller selectivelycoupling the output of the first mixer to the second bandpass filter forDMD phase noise measurement, and the controller selectively coupling theoutput of the first mixer to the first low pass filter for additivephase noise measurement; a second mixer having a first input, a secondinput, and an output, an output of the first bandpass filter beingcoupled to the first input of the second mixer; a phase locked loopcoupled to a second phase shifter, an output of the first bandpassfilter being coupled to the phase locked loop, an output of the secondphase shifter being coupled to the second input of the second mixer; asecond low pass filter having an input coupled to the output of thesecond mixer; and a comb generator coupled to an output of the secondbandpass filter, the controller selectively coupling the outputs of thefirst low pass filter, the second lowpass filter, and the combgenerator, to the output of the receiver/downconverter.
 7. The phasenoise measurement system according to claim 6, wherein the low noisesignal synthesizer further includes:a low noise crystal oscillator forproducing a signal having a first frequency; a surface acoustic waveoscillator for producing a signal having a second frequency; a combgenerator having an input and an output, the input of the comb generatorcoupled to an output of the crystal oscillator; a bandpass filter havingan output and an input, the input of the bandpass filter being coupledto the output of the comb generator, the bandpass filter having apassband which includes the second frequency; a frequency dividingcomponent coupled to the low noise crystal oscillator, the controllercontrolling the frequency dividing component to selectively produce oneof a plurality of offset frequencies, each of the plurality of offsetfrequencies being in a first frequency range; a mixer having a firstinput and a second input, the first input coupled to an output of thefrequency dividing component, the controller selectively coupling thesecond input to one of the output of the surface acoustic waveoscillator and the output of the bandpass filter; and a tunable bandpassfilter coupled to an output of the mixer, the controller selectivelytuning the tunable bandpass filter to a passband which includes a sum ofthe selected offset frequency and the second frequency.
 8. The phasenoise measurement system according to claim 7, wherein the secondfrequency is 960 MHz.
 9. The phase noise measurement system according toclaim 7, wherein the first frequency is 120 MHz.
 10. The phase noisemeasurement system according to claim 7, wherein the first frequencyrange is from 10 MHz to 40 MHZ.
 11. The phase noise measurement systemaccording to claim 7, wherein the passband of the bandpass filter iscentered at 960 MHz.
 12. The phase noise measurement system according toclaim 7, wherein the controller selectively tunes the tunable bandpassfilter to a difference between the selected offset frequency and thesecond frequency.
 13. The phase noise measurement system according toclaim 6, further comprising a delay element, the delay element beingcoupled between the input of the first phase shifter and the UUT inputfor additive phase noise measurement.
 14. The phase noise measurementsystem according to claim 6, wherein the passband of the first bandpassfilter is centered at 100 MHz.
 15. The phase noise measurement systemaccording to claim 6, wherein the passband of the second bandpass filteris centered at 10 MHz.
 16. The phase noise measurement system accordingto claim 6, wherein the cutoff frequency of the first and second lowpassfilters is 1.9 MHz.